ADVANTAGES OF FSK OVER AM.—Frequency-shift keying is used in all single-channel, radiotelegraph systems that use automatic printing systems. The advantage of fsk over on-off keyed cw is that it rejects unwanted signals (noise) that are weaker than the desired signal. This is true of all fm systems. Also, since a signal is always present in the fsk receiver, automatic volume control methods may be used to minimize the effects of signal fading caused by ionospheric variations. The amount of inherent signal-to-noise ratio improvement of fsk over AM is approximately 3 to 4 dB. This improvement is because the signal energy of fsk is always present while signal energy is present for only one-half the time in AM systems. Noise is continuously present in both fsk and AM, but is eliminated in fsk reception. Under the rapid fading and high-noise conditions that commonly exist in the high frequency (hf) region, fsk shows a marked advantage over AM. Overall improvement is sometimes expressed as the RATIO OF TRANSMITTED POWERS required to give equivalent transmission results over the two systems. Such a ratio varies widely, depending on the prevailing conditions. With little fading, the ratio may be entirely the result of the improvement in signal-to-noise ratio and may be under 5 dB. However, under severe fading conditions, large amounts of power often fail to give good results for AM transmission. At the same time, fsk may be satisfactory at nominal power. The power ratio (fsk versus AM) would become infinite in such a case.
Another application of fsk is at low and very low frequencies (below 300 kilohertz). At these frequencies, keying speeds are limited by the "flywheel" effect of the extremely large capacitance and inductance of the antenna circuits. These circuits tend to oscillate at their resonant frequencies. Frequency-shifting the transmitter and changing the antenna resonance by the same keying impulses will result in much greater keying speeds. As a result, the use of these expensive channels is much more efficient.
Q-1. What are the two types of angle modulation? Q-2. Name the modulation system in which the frequency alternates between two discrete values in
response to the opening and closing of a key? Q-3. What is the primary advantage of an fsk transmission system?
In frequency modulation, the instantaneous frequency of the radio-frequency wave is varied in accordance with the modulating signal, as shown in view (A) of figure 2-5. As mentioned earlier, the amplitude is kept constant. This results in oscillations similar to those illustrated in view (B). The number
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of times per second that the instantaneous frequency is varied from the average (carrier frequency) is controlled by the frequency of the modulating signal. The amount by which the frequency departs from the average is controlled by the amplitude of the modulating signal. This variation is referred to as the FREQUENCY DEVIATION of the frequency-modulated wave. We can now establish two clear-cut rules for frequency deviation rate and amplitude in frequency modulation:
(This rule simply means that if a 10-volt signal causes a frequency shift of 20 kilohertz, then a 20-volt signal will cause a frequency shift of 40 kilohertz.)
(This second rule means that if the carrier is modulated with a 1-kilohertz tone, then the carrier is changing frequency 1,000 times each second.)
Figure 2-6 illustrates a simple oscillator circuit with the addition of a condenser microphone (M) in shunt with the oscillator tank circuit. Although the condenser microphone capacitance is actually very low, the capacitance of this microphone will be considered near that of the tuning capacitor (C). The frequency of oscillation in this circuit is, of course, determined by the LC product of all elements of the circuit; but, the product of the inductance (L) and the combined capacitance of C and M are the primary frequency components. When no sound waves strike M, the frequency is the rf carrier frequency. Any excitation of M will alter its capacitance and, therefore, the frequency of the oscillator circuit. Figure 2-7 illustrates what happens to the capacitance of the microphone during excitation. In view (A), the audio- frequency wave has three levels of intensity, shown as X, a whisper; Y, a normal voice; and Z, a loud voice. In view (B), the same conditions of intensity are repeated, but this time at a frequency twice that of view (A). Note in each case that the capacitance changes both positively and negatively; thus the frequency of oscillation alternates both above and below the resting frequency. The amount of change is determined by the change in capacitance of the microphone. The change is caused by the amplitude of the sound wave exciting the microphone. The rate at which the change in frequency occurs is determined by
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the rate at which the capacitance of the microphone changes. This rate of change is caused by the frequency of the sound wave. For example, suppose a 1,000-hertz tone of a certain loudness strikes the microphone. The frequency of the carrier will then shift by a certain amount, say plus and minus 40 kilohertz. The carrier will be shifted 1,000 times per second. Now assume that with its loudness unchanged, the frequency of the tone is changed to 4,000 hertz. The carrier frequency will still shift plus and minus 40 kilohertz; but now it will shift at a rate of 4,000 times per second. Likewise, assume that at the same loudness, the tone is reduced to 200 hertz. The carrier will continue to shift plus and minus 40 kilohertz, but now at a rate of 200 times per second. If the loudness of any of these modulating tones is reduced by one-half, the frequency of the carrier will be shifted plus and minus 20 kilohertz. The carrier will then shift at the same rate as before. This fulfills all requirements for frequency modulation. Both the frequency and the amplitude of the modulating signal are translated into variations in the frequency of the rf carrier.
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Figure 2-8 shows how the frequency shift of an fm signal goes through the same variations as does the modulating signal. In this figure the dimension of the constant amplitude is omitted. (As these remaining waveforms are presented, be sure you take plenty of time to study and digest what the figures tell you. Look each one over carefully, noting everything you can about them. Doing this will help you understand this material.) If the maximum frequency deviation is set at 75 kilohertz above and below the carrier, the audio amplitude of the modulating wave must be so adjusted that its peaks drive the frequency only between these limits. This can then be referred to as 100-PERCENT MODULATION, although the term is only remotely applicable to fm. Projections along the vertical axis represent deviations in frequency from the resting frequency (carrier) in terms of audio amplitude. Projections along the horizontal axis represent time. The distance between A and B represents 0.001 second. This means that carrier deviations from the resting frequency to plus 75 kilohertz, then to minus 75 kilohertz, and finally back to rest would occur 1,000 times per second. This would equate to an audio frequency of 1,000 hertz. Since the carrier deviation for this period (A to B) extends to the full allowable limits of plus and minus 75 kilohertz, the wave is fully modulated. The distance from C to D is the same as that from A to B, so the time interval and frequency are the same as before. Notice, however, that the amplitude of the modulating wave has been decreased so that the carrier is driven to only plus and minus 37.5 kilohertz, one-half the allowable deviation. This would correspond to only 50-percent modulation if the system were AM instead of fm. Between E and F, the interval is reduced to 0.0005 second. This indicates an increase in frequency of the modulating signal to 2,000 hertz. The amplitude has returned to its maximum allowable value, as indicated by the deviation of the carrier to plus and minus 75 kilohertz. Interval G to H represents the same frequency at a lower modulation amplitude (66 percent). Notice the GUARD BANDS between plus and minus 75 kilohertz and plus and minus 100 kilohertz. These bands isolate the modulation extremes of this particular channel from that of adjacent channels.
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PERCENT OF MODULATION.—Before we explain 100-percent modulation in an fm system, let's review the conditions for 100-percent modulation of an AM wave. Recall that 100-percent modulation for AM exists when the amplitude of the modulation envelope varies between 0 volts and twice its normal unmodulated value. At 100-percent modulation there is a power increase of 50 percent. Because the modulating wave is not constant in voice signals, the degree of modulation constantly varies. In this case the vacuum tubes in an AM system cannot be operated at maximum efficiency because of varying power requirements.
In frequency modulation, 100-percent modulation has a meaning different from that of AM. The modulating signal varies only the frequency of the carrier. Therefore, tubes do not have varying power requirements and can be operated at maximum efficiency and the fm signal has a constant power output. In fm a modulation of 100 percent simply means that the carrier is deviated in frequency by the full permissible amount. For example, an 88.5-megahertz fm station operates at 100-percent modulation when the modulating signal deviation frequency band is from 75 kilohertz above to 75 kilohertz below the carrier (the maximum allowable limits). This maximum deviation frequency is set arbitrarily and will vary according to the applications of a given fm transmitter. In the case given above, 50-percent modulation would mean that the carrier was deviated 37.5 kilohertz above and below the resting frequency (50 percent of the 150-kilohertz band divided by 2). Other assignments for fm service may limit the allowable deviation to 50 kilohertz, or even 10 kilohertz. Since there is no fixed value for comparison, the term "percent of modulation" has little meaning for fm. The term MODULATION INDEX is more useful in fm modulation discussions. Modulation index is frequency deviation divided by the frequency of the modulating signal.
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MODULATION INDEX.—This ratio of frequency deviation to frequency of the modulating signal is useful because it also describes the ratio of amplitude to tone for the audio signal. These factors determine the number and spacing of the side frequencies of the transmitted signal. The modulation index formula is shown below:
Views (A) and (B) of figure 2-9 show the frequency spectrum for various fm signals. In the four examples of view (A), the modulating frequency is constant; the deviation frequency is changed to show the effects of modulation indexes of 0.5, 1.0, 5.0, and 10.0. In view (B) the deviation frequency is held constant and the modulating frequency is varied to give the same modulation indexes.
You can determine several facts about fm signals by studying the frequency spectrum. For example, table 2-1 was developed from the information in figure 2-9. Notice in the top spectrums of both views (A) and (B) that the modulation index is 0.5. Also notice as you look at the next lower spectrums that the modulation index is 1.0. Next down is 5.0, and finally, the bottom spectrums have modulation indexes of
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10.0. This information was used to develop table 2-1 by listing the modulation indexes in the left column and the number of significant sidebands in the right. SIGNIFICANT SIDEBANDS (those with significantly large amplitudes) are shown in both views of figure 2-9 as vertical lines on each side of the carrier frequency. Actually, an infinite number of sidebands are produced, but only a small portion of them are of sufficient amplitude to be important. For example, for a modulation index of 0.5 [top spectrums of both views (A) and (B)], the number of significant sidebands counted is 4. For the next spectrums down, the modulation index is 1.0 and the number of sidebands is 6, and so forth. This holds true for any combination of deviating and modulating frequencies that yield identical modulating indexes.
Table 2-1.—Modulation index table
.01 2 .4 2 .5 4
1.0 6 2.0 8 3.0 12 4.0 14 5.0 16 6.0 18 7.0 22 8.0 24 9.0 26
10.0 28 11.0 32 12.0 32 13.0 36 14.0 38 15.0 38
You should be able to see by studying figure 2-9, views (A) and (B), that the modulating frequency determines the spacing of the sideband frequencies. By using a significant sidebands table (such as table 2-1), you can determine the bandwidth of a given fm signal. Figure 2-10 illustrates the use of this table. The carrier frequency shown is 500 kilohertz. The modulating frequency is 15 kilohertz and the deviation frequency is 75 kilohertz.
2-15
77
From table 2-1 we see that there are 16 significant sidebands for a modulation index of 5. To determine total bandwidth for this case, we use:
The use of this math is to illustrate that the actual bandwidth of an fm transmitter (240 kHz) is greater than that suggested by its maximum deviation bandwidth ( – 75 kHz or 150 kHz). This is important to know when choosing operating frequencies or designing equipment.
Q-4. What characteristic of a carrier wave is varied in frequency modulation? Q-5. How is the degree of modulation expressed in an fm system? Q-6. What two values may be used to determine the bandwidth of an fm wave?
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METHODS OF FREQUENCY MODULATION.—The circuit shown earlier in figure 2-6 and the discussion in previous paragraphs were for illustrative purposes only. In reality, such a circuit would not be practical. However, the basic principle involved (the change in reactance of an oscillator circuit in accordance with the modulating voltage) constitutes one of the methods of developing a frequency- modulated wave.
Reactance-Tube Modulation.—In direct modulation, an oscillator is frequency modulated by a REACTANCE TUBE that is in parallel (SHUNT) with the oscillator tank circuit. (The terms "shunt" or "shunting" will be used in this module to mean the same as "parallel" or "to place in parallel with" components.) This is illustrated in figure 2-11. The oscillator is a conventional Hartley circuit with the reactance-tube circuit in parallel with the tank circuit of the oscillator tube. The reactance tube is an ordinary pentode. It is made to act either capacitively or inductively; that is, its grid is excited with a voltage which either leads or lags the oscillator voltage by 90 degrees.
When the reactance tube is connected across the tank circuit with no modulating voltage applied, it will affect the frequency of the oscillator. The voltage across the oscillator tank circuit (L1 and C1) is also in parallel with the series network of R1 and C7. This voltage causes a current flow through R1 and C7. If R1 is at least five times larger than the capacitive reactance of C7, this branch of the circuit will be essentially resistive. Voltage E
1, which is across C7, will lag current by 90 degrees. E 1 is applied to the control grid of reactance tube V1. This changes plate current (Ip), which essentially flows only through the LC tank circuit. This is because the value of R1 is high compared to the impedance of the tank circuit. Since current is inversely proportional to impedance, most of the plate current coupled through C3 flows through the tank circuit.
At resonance, the voltage and current in the tank circuit are in phase. Because E1 lags E by 90 degrees and I
is in phase with grid voltage E1, the superimposed current through the tank circuit lags the original tank current by 90 degrees. Both the resultant current (caused by Ip) and the tank current lag tank voltage and current by some angle depending on the relative amplitudes of the two currents. Because this resultant current is a lagging current, the impedance across the tank circuit cannot be at its maximum unless something happens within the tank to bring current and voltage into phase. Therefore, this situation continues until the frequency of oscillations in the tank circuit changes sufficiently so that the voltages across the tank and the current flowing into it are again in phase. This action is the same as would be produced by adding a reactance in parallel with the L1C1 tank. Because the superimposed current lags voltage E by 90 degrees, the introduced reactance is inductive. In NEETS, Module 2, Introduction to
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Alternating Current and Transformers, you learned that total inductance decreases as additional inductors are added in parallel. Because this introduced reactance effectively reduces inductance, the frequency of the oscillator increases to a new fixed value.
Now let’s see what happens when a modulating signal is applied. The magnitude of the introduced reactance is determined by the magnitude of the superimposed current through the tank. The magnitude of Ip for a given E1 is determined by the transconductance of V1. (Transconductance was covered in NEETS, Module 6, Introduction to Electronic Emission, Tubes, and Power Supplies.) Therefore, the value of reactance introduced into the tuned circuit varies directly with the transconductance of the reactance tube. When a modulating signal is applied to the grid of V1, both E1 and I change, causing transconductance to vary with the modulating signal. This causes a variable reactance to be introduced into the tuned circuit. This variable reactance either adds to or subtracts from the fixed value of reactance that is introduced in the absence of the modulating signal. This action varies the reactance across the oscillator which, in turn, varies the instantaneous frequency of the oscillator. These variations in the oscillator frequency are proportional to the instantaneous amplitude of the modulating voltage. Reactance-tube modulators are usually operated at low power levels. The required output power is developed in power amplifier stages that follow the modulators.
The output of a reactance-tube modulated oscillator also contains some unwanted amplitude modulation. This unwanted modulation is caused by stray capacitance and the resistive component of the RC phase splitter. The resistance is much less significant than the desired XC, but the resistance does allow some plate current to flow which is not of the proper phase relationship for good tube operation. The small amplitude modulation that this produces is easily removed by passing the oscillator output through a limiter-amplifier circuit.
Semiconductor Reactance Modulator.—The SEMICONDUCTOR-REACTANCE MODULATOR is used to frequency modulate low-power semiconductor transmitters. Figure 2-12 shows a typical frequency-modulated oscillator stage operated as a reactance modulator. Q1, along with its associated circuitry, is the oscillator. Q2 is the modulator and is connected to the circuit so that its collector-to- emitter capacitance (CCE) is in parallel with a portion of the rf oscillator coil, L1. As the modulator operates, the output capacitance of Q2 is varied. Thus, the frequency of the oscillator is shifted in accordance with the modulation the same as if C1 were varied.
2-18
When the modulating signal is applied to the base of Q2, the emitter-to-base bias varies at the modulation rate. This causes the collector voltage of Q2 to vary at the same modulating rate. When the collector voltage increases, output capacitance CCE decreases; when the collector voltage decreases, CCE increases. An increase in collector voltage has the effect of spreading the plates of CCE farther apart by increasing the width of the barrier. A decrease of collector voltage reduces the width of the pn junction and has the same effect as pushing the capacitor plates together to provide more capacitance.
When the output capacitance decreases, the instantaneous frequency of the oscillator tank circuit increases (acts the same as if C1 were decreased). When the output capacitance increases, the instantaneous frequency of the oscillator tank circuit decreases. This decrease in frequency produces a lower frequency in the output because of the shunting effect of CCE. Thus, the frequency of the oscillator tank circuit increases and decreases at an audio frequency (af) modulating rate. The output of the oscillator, therefore, is a frequency modulated rf signal.
Since the audio modulation causes the collector voltage to increase and decrease, an AM component is induced into the output. This produces both an fm and AM output. The amplitude variations are then removed by placing a limiter stage after the reactance modulator and only the frequency modulation remains.
Frequency multipliers or mixers (discussed in chapter 1) are used to increase the oscillator frequency to the desired output frequency. For high-power applications, linear rf amplifiers are used to increase the steady-amplitude signal to a higher power output. With the initial modulation occurring at low levels, fm represents a savings of power when compared to conventional AM. This is because fm noise-reducing properties provide a better signal-to-noise ratio than is possible with AM.
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Multivibrator Modulator.—Another type of frequency modulator is the astable multivibrator illustrated in figure 2-13. Inserting the modulating af voltage in series with the base-return of the multivibrator transistors causes the gate length, and thus the fundamental frequency of the multivibrator, to vary. The amount of variation will be in accordance with the amplitude of the modulating voltage. One requirement of this method is that the fundamental frequency of the multivibrator be high in relation to the highest modulating frequencies. A factor of at least 100 provides the best results.
Recall that a multivibrator output consists of the fundamental frequency and all of its harmonics. Unwanted even harmonics are eliminated by using a SYMMETRICAL MULTIVIBRATOR circuit, as shown in figure 2-13. The desired fundamental frequency, or desired odd harmonics, can be amplified after all other odd harmonics are eliminated in the LCR filter section of figure 2-13. A single frequency- modulated carrier is then made available for further amplification and transmission.
Proper design of the multivibrator will cause the frequency deviation of the carrier to faithfully follow (referred to as a "linear" function) the modulating voltage. This is true up to frequency deviations which are considerable fractions of the fundamental frequency of the multivibrator. The principal design consideration is that the RC coupling from one multivibrator transistor base to the collector of the other has a time constant which is greater than the actual gate length by a factor of 10 or more. Under these conditions, a rise in base voltage in each transistor is essentially linear from cutoff to the bias at which the transistor is switched on. Since this rise in base voltage is a linear function of time, the gate length will change as an inverse function of the modulating voltage. This action will cause the frequency to change as a linear function of the modulating voltage.
The multivibrator frequency modulator has the advantage over the reactance-type modulator of a greater linear frequency deviation from a given carrier frequency. However, multivibrators are limited to frequencies below about 1 megahertz. Both systems are subject to drift of the carrier frequency and must, therefore, be stabilized. Stabilization may be accomplished by modulating at a relatively low frequency and translating by heterodyne action to the desired output frequency, as shown in figure 2-14. A 1-megahertz signal is heterodyned with 49 megahertz from the crystal-controlled oscillator to provide a stable 50-megahertz output from the mixer. If a suitably stable heterodyning oscillator is used, the frequency stability can be greatly improved. For instance, at the frequencies shown in figure 2-14, the
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stability of the unmodulated 50-megahertz carrier would be 50 times better than that which harmonic multiplication could provide.
Varactor FM Modulator.—Another fm modulator which is widely used in transistorized circuitry uses a voltage-variable capacitor (VARACTOR). The varactor is simply a diode, or pn junction, that is designed to have a certain amount of capacitance between junctions. View (A) of figure 2-15 shows the varactor schematic symbol. A diagram of a varactor in a simple oscillator circuit is shown in view (B). This is not a working circuit, but merely a simplified illustration. The capacitance of a varactor, as with regular capacitors, is determined by the area of the capacitor plates and the distance between the plates. The depletion region in the varactor is the dielectric and is located between the p and n elements, which serve as the plates. Capacitance is varied in the varactor by varying the reverse bias which controls the thickness of the depletion region. The varactor is so designed that the change in capacitance is linear with the change in the applied voltage. This is a special design characteristic of the varactor diode. The varactor must not be forward biased because it cannot tolerate much current flow. Proper circuit design prevents the application of forward bias.
2-21
Notice the simplicity of operation of the circuit in figure 2-16. An af signal that is applied to the input results in the following actions: (1) On the positive alternation, reverse bias increases and the dielectric (depletion region) width increases. This decreases capacitance which increases the frequency of the oscillator. (2) On the negative alternation, the reverse bias decreases, which results in a decrease in oscillator frequency.
Many different fm modulators are available, but they all use the basic principles you have just studied. The main point to remember is that an oscillator must be used to establish the reference (carrier) frequency. Secondly, some method is needed to cause the oscillator to change frequency in accordance with an af signal. Anytime this can be accomplished, we have a frequency modulator.
Q-7. How does the reactance-tube modulator impress intelligence onto an rf carrier? Q-8. What characteristic of a transistor is varied in a semiconductor-reactance modulator? Q-9. What circuit section is required in the output of a multivibrator modulator to eliminate
unwanted output frequencies? Q-10. What characteristic of a varactor is used in an fm modulator?
2-22
Frequency modulation requires the oscillator frequency to deviate both above and below the carrier frequency. During the process of frequency modulation, the peaks of each successive cycle in the modulated waveform occur at times other than they would if the carrier were unmodulated. This is actually an incidental phase shift that takes place along with the frequency shift in fm. Just the opposite action takes place in phase modulation. The af signal is applied to a PHASE MODULATOR in pm. The resultant wave from the phase modulator shifts in phase, as illustrated in figure 2-17. Notice that the time period of each successive cycle varies in the modulated wave according to the audio-wave variation. Since frequency is a function of time period per cycle, we can see that such a phase shift in the carrier will cause its frequency to change. The frequency change in fm is vital, but in pm it is merely incidental. The amount of frequency change has nothing to do with the resultant modulated wave shape in pm. At this point the comparison of fm to pm may seem a little hazy, but it will clear up as we progress.
Let’s review some voltage phase relationships. Look at figure 2-18 and compare the three voltages (A, B, and C). Since voltage A begins its cycle and reaches its peak before voltage B, it is said to lead voltage B. Voltage C, on the other hand, lags voltage B by 30 degrees. In phase modulation the phase of the carrier is caused to shift at the rate of the af modulating signal. In figure 2-19, note that the unmodulated carrier has constant phase, amplitude, and frequency. The dotted wave shape represents the modulated carrier. Notice that the phase on the second peak leads the phase of the unmodulated carrier. On the third peak the shift is even greater; however, on-the fourth peak, the peaks begin to realign phase with each other. These relationships represent the effect of 1/2 cycle of an af modulating signal. On the negative alternation of the af intelligence, the phase of the carrier would lag and the peaks would occur at times later than they would in the unmodulated carrier.
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The presentation of these two waves together does not mean that we transmit a modulated wave together with an unmodulated carrier. The two waveforms were drawn together only to show how a modulated wave looks when compared to an unmodulated wave.
Now that you have seen the phase and frequency shifts in both fm and pm, let’s find out exactly how they differ. First, only the phase shift is important in pm. It is proportional to the af modulating signal. To visualize this relationship, refer to the wave shapes shown in figure 2-20. Study the composition of the fm and pm waves carefully as they are modulated with the modulating wave shape. Notice that in fm, the carrier frequency deviates when the modulating wave changes polarity. With each alternation of the modulating wave, the carrier advances or retards in frequency and remains at the new frequency for the duration of that cycle. In pm you can see that between one alternation and the next, the carrier phase must change, and the frequency shift that occurs does so only during the transition time; the frequency then returns to its normal rate. Note in the pm wave that the frequency shift occurs only when the modulating wave is changing polarity. The frequency during the constant amplitude portion of each alternation is the REST FREQUENCY.
The relationship, in pm, of the modulating af to the change in the phase shift is easy to see once you understand AM and fm principles. Again, we can establish two clear-cut rules of phase modulation:
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• AMOUNT OF PHASE SHIFT IS PROPORTIONAL TO THE AMPLITUDE OF THE MODULATING SIGNAL.
(If a 10-volt signal causes a phase shift of 20 degrees, then a 20-volt signal causes a phase shift of 40 degrees.)
• RATE OF PHASE SHIFT IS PROPORTIONAL TO THE FREQUENCY OF THE MODULATING SIGNAL.
(If the carrier were modulated with a 1-kilohertz tone, the carrier would advance and retard in phase 1,000 times each second.)
Phase modulation is also similar to frequency modulation in the number of sidebands that exist within the modulated wave and the spacing between sidebands. Phase modulation will also produce an infinite number of sideband frequencies. The spacing between these sidebands will be equal to the frequency of the modulating signal. However, one factor is very different in phase modulation; that is, the distribution of power in pm sidebands is not similar to that in fm sidebands, as will be explained in the next section.
Recall from frequency modulation that modulation index is used to calculate the number of significant sidebands existing in the waveform. The higher the modulation index, the greater the number of sideband pairs. The modulation index is the ratio between the amount of oscillator deviation and the frequency of the modulating signal:
In frequency modulation, we saw that as the frequency of the modulating signal increased (assuming the deviation remained constant) the number of significant sideband pairs decreased. This is shown in views (A) and (B) of figure 2-21. Notice that although the total number of significant sidebands decreases with a higher frequency-modulating signal, the sidebands spread out relative to each other; the total bandwidth increases.
2-25
In phase modulation the oscillator does not deviate, and the power in the sidebands is a function of the amplitude of the modulating signal. Therefore, two signals, one at 5 kilohertz and the other at 10 kilohertz, used to modulate a carrier would have the same sideband power distribution. However, the 10-kilohertz sidebands would be farther apart, as shown in views (C) and (D) of figure 2-21. When compared to fm, the bandwidth of the pm transmitted signal is greatly increased as the frequency of the modulating signal is increased.
As we pointed out earlier, phase modulation cannot occur without an incidental change in frequency, nor can frequency modulation occur without an incidental change in phase. The term fm is loosely used when referring to any type of angle modulation, and phase modulation is sometimes incorrectly referred to as "indirect fm." This is a definition that you should disregard to avoid confusion. Phase modulation is just what the words imply — phase modulation of a carrier by an af modulating signal. You will develop a better understanding of these points as you advance in your study of modulation.
In phase modulation you learned that varying the phase of a carrier at an intelligence rate caused that carrier to contain variations which could be converted back into intelligence. One circuit that can cause this phase variation is shown in figure 2-22.
2-26
The capacitor in series with the resistor forms a phase-shift circuit. With a constant frequency rf carrier applied at the input, the output across the resistor would be 45 degrees out of phase with the input if X
C = R.
Now, let’s vary the resistance and observe how the output is affected in figure 2-23. As the resistance reaches a value greater than 10 times XC, the phase difference between input and output is nearly 0 degrees. For all practical purposes, the circuit is resistive. As the resistance is decreased to 1/10 the value of X
C, the phase difference approaches 90 degrees. The circuit is now almost completely capacitive. By replacing the resistor with a vacuum tube, as shown in view (A) of figure 2-24, we can vary the resistance (vacuum-tube impedance) by varying the voltage applied to the grid of the tube. The frequency applied to the circuit (from a crystal-controlled master oscillator) will be shifted in phase by 45 degrees with no audio input [view (B)]. With the application of an audio signal, the phase will shift as the impedance of the tube is varied.
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In practice, a circuit like this could not provide enough phase shift to produce the desired results in the output. Several of these circuits are arranged in cascade to provide the desired amount of phase shift. Also, since the output of this circuit will vary in amplitude, the signal is fed to a limiter to remove amplitude variations.
The major advantage of this type modulation circuit over frequency modulation is that this circuit uses a crystal-controlled oscillator to maintain a stable carrier frequency. In fm the oscillator cannot be crystal controlled because it is actually required to vary in frequency. That means that an fm oscillator will require a complex automatic frequency control (afc) system. An afc system ensures that the oscillator stays on the same carrier frequency and achieves a high degree of stability. The afc circuit will be covered in a later module.
Phase-shift keying (psk) is similar to ON-OFF cw keying in AM systems and frequency-shift keying in fm systems. Psk is most useful when the code elements are all of equal length; that is, all marks and spaces, whether message elements or synchronizing signals, occupy identical elements of time. It is not fully suitable for use on start-stop teletypewriter circuits where the stop pulse is 1.42 times longer than the
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other pulses. Neither is it applicable to those pulsed systems in which the duration or position of the pulses are varied by the modulation frequency. In its simplest form, psk operates on the principle of phase reversal of the carrier. Each time a mark is received, the phase is reversed. No phase reversal takes place when a space is received. In binary systems, marks and spaces are called ONES and ZEROS, respectively, so that a ONE causes a 180-degree phase shift, and a ZERO has no effect on the incoming signal. Figure 2-25 shows the application of phase-shift keying to an unmodulated carrier [view (A)] in the af range. For transmission over other than a conductive path, the wave shown in view (D) must be used as the modulating signal for some other system of modulating an rf carrier.
The modulating signal in view (B) consists of a bit stream of ZEROS and ONES. A ZERO does not affect the carrier frequency which is usually set to equal the bit rate. For example, a data stream of 1,200 bits per second would have a carrier of 1,200 hertz. When a data bit ONE occurs, the phase of the carrier frequency is shifted 180 degrees. In view (C) we find that the third, fifth, and sixth cycles (all ONE) have been reversed in phase. This phase reversal produces CUSPS (sharp phase reversals) which are usually
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removed by filtering before transmission or further modulation. This filtering action limits the bandwidth of the output signal frequencies. The resulting wave is shown in view (D).
The exact waveform of figure 2-25, view (D), can be obtained by logic operations of timing and data. This is illustrated in figure 2-26, where a timing signal [view (A)] is used rather than a carrier frequency. The data (intelligence) is shown in view (B) and is combined with the timing signal to produce a combination digital modulation signal, as shown in view (C). The square-wave pattern of the digital modulation is filtered to limit the bandwidth of the signal frequencies, as shown in view (D). This system has been used in some high-speed data equipment, but it offers no particular advantage over other systems of modulation, particularly the pulse-modulated systems for high-speed data transmission.
Q-11. What type of modulation depends on the carrier-wave phase shift? Q-12. What components may be used to build a basic phase modulator? Q-13. Phase-shift keying is similar to what other two types of modulation?
2-30
Another type of modulation is PULSE MODULATION. Pulse modulation has many uses, including telegraphy, radar, telemetry, and multiplexing. Far too many applications of pulse modulation exist to elaborate on any one of them, but in this section we will cover the basic principles of pulse modulation.
Amplitude modulating a simple rf carrier to a point where it becomes drastically overmodulated could produce a waveform similar to that required in pulse modulation. A modulating signal [view (A) of figure 2-27] that is much larger than the carrier results in the modulation envelope shown in view (B). The modulation envelope would be the same if the modulating wave shape were not sinusoidal; that is, like the one shown in view (C).
2-31
Observe the modulating square wave in figure 2-28. Remember that it contains an infinite number of odd harmonics in addition to its fundamental frequency. Assume that a carrier has a frequency of 1 megahertz. The fundamental frequency of the modulating square wave is 1 kilohertz. When these signals heterodyne, two new frequencies will be produced: a sum frequency of 1.001 megahertz and a difference frequency of 0.999 megahertz. The fundamental frequency heterodynes with the carrier. This is also true of all harmonics contained in the square wave. Side frequencies associated with those harmonics will be produced as a result of this process. For example, the third harmonic of the square wave heterodynes with the carrier and produces sideband frequencies at 1.003 and 0.997 megahertz. Another set will be produced by the fifth, seventh, ninth, eleventh, thirteenth, fifteenth, seventeenth, and nineteenth harmonics of the square wave, and so on to infinity.
Look at figure 2-28 and observe the relative amplitudes of the sidebands as they relate to the amplitudes of the harmonics contained in the square wave. Note that the first set of sidebands is directly related to the amplitude of the square wave. The second set of sidebands is related to the third harmonic content of the square wave and is 1/3 the amplitude of the first set. The third set is related to the amplitude of the first set of sidebands and is 115 the amplitude of the first set. This relationship will apply to each additional set of sidebands.
2-32
View (A) of figure 2-29 shows the carrier modulated with a square wave. In view (B) the modulating square wave is increased in amplitude; note that the rf peaks increase in amplitude during the positive alternation of the square wave and decrease during the negative half of the square wave. In view (C) the amplitude of the square wave is further increased and the amplitude of the rf wave is almost 0 during the negative alternation of the square wave.
2-33
Note the frequency spectrum associated with each of these conditions. The carrier amplitude remains constant, but the sidebands increase in amplitude in accordance with the amplitude of the modulating square wave.
So far in pulse modulation, the same general rules apply as in AM. In view (C) the amplitude of the square wave of voltage is equal to the peak voltage of the unmodulated carrier wave. This is 100-percent modulation, just as in conventional AM. Note in the frequency spectrum that the sideband distribution is also the same as in AM. Keep in mind that the total sideband power is 1/2 of the total power when the modulator signal is a square wave. This is in contrast to 1/3 the total power with sine-wave modulation.
Now refer to view (D). The increase of the square-wave modulating voltage is greater in amplitude than the unmodulated carrier. Notice that the sideband distribution does not change; but, as the sidebands take on more of the transmitted power, so will the carrier.
Thus far, we have established a carrier and have caused its peaks to increase and decrease as a modulating square wave is applied. Some pulse-modulation systems modulate a carrier in this manner. Others produce no rf until pulsed; that is, rf occurs only during the actual pulse as shown in view (A) of figure 2-30. For example, let’s start with an rf carrier frequency of 1 megahertz. Each cycle of the rf requires a certain amount of time to complete. If we allow oscillations to occur for a given period of time only during selected intervals, as in view (B), we are PULSING the system. Note that the pulse transmitter does not produce an rf signal until one of the positive-going modulating pulses is applied. The transmitter then produces the rf carrier until the positive input pulse ends and the input waveform again becomes a negative potential.
Refer back to figure 1-41 and the over-modulation discussion in chapter 1. You will notice that the overmodulation wave shape of view (D) in figure 2-29 and the pulse-modulation wave shape of figure 2-30, view (B), are very similar to figure 1-41.
Actually, both figure 1-41 and view (D) of figure 2-29 result from overmodulation. Even though the output of the pulse transmitter in figure 2-30 looks like overmodulation, it is not; rather, it is pulsed.
2-34
However, the frequency spectrums are similar. Sideband distributions are similar, but not identical, since the pulse transmitter in figure 2-30 is gated on and off instead of being modulated by a square wave as was the case in view (D) of figure 2-29.
Remember, in pulse modulation the sidebands produced to accompany the carrier during transmission are directly related to the harmonic content of the modulating wave shape. In figure 2-31, (view A, view B and view C), observe the square and rectangular wave shapes used to pulse modulate the same carrier frequency in each of the three views.
Let’s take note of some timing relationships in the three modulating sequences in figure 2-31:
2-35
In figure 2-32, observe the relative time for individual rf cycles. The time for each cycle is the same in views (A) and (B). Since this time is the same, we can assume that the carrier frequency is the same. But in view (C) the time for each cycle is about half that in views (A) and (B). Therefore, the frequency of the carrier in view (C) is nearly twice that of the other two. This illustration shows that carrier frequencies in pulse systems can vary.
The carrier frequency is not the only frequency we must concern ourselves with in pulse systems. We must also be concerned with the frequency that is associated with the repetition rate of groups of pulses. Figure 2-33 shows that a specific time period exists between each group of rf pulses. This time is the same for each repetition of the pulse and is called the PULSE-REPETITION TIME (prt). To find out how often these groups of pulses occur, compute PULSE-REPETITION FREQUENCY (prf) using the formula:
2-36
Just remember that the pulse-repetition time is the time it takes for a pulse to recur, as shown in figure 2-34. The duration of time of the pulse (a) plus the time when no pulse occurs (b) equals the total pulse-repetition time.
The time during which the pulse is occurring is called PULSE DURATION (pd) or PULSE WIDTH (pw), as shown in figure 2-35. As you will soon see, pulse width is important in pulse modulation.
2-37
The time we have been referring to as the time of no pulse, or nonpulse time, is referred to as REST TIME (rt). The duration of this rest time will determine certain capabilities of the pulse-modulation system. The pulse width is the time that the transmitter produces rf oscillations and is the actual pulse transmission time. During the nonpulse time, shown in figure 2-36, the transmitter produces no oscillations and the oscillator is cut off.
Some pulse transmitter-receiver systems transmit the pulse and then rest, awaiting the return of an echo. Rest time provides the system time for the receive cycle of operation.
When discussing power in a pulse-modulation system, we have to consider PEAK POWER and AVERAGE POWER. Peak power is the maximum value of the transmitted pulse; average power is the peak power value averaged over the pulse-repetition time. Peak power is very easy to see in a pulse system. In figure 2-37, all pulsed wave shapes have a peak power of 100 watts. Also note that in views (A), (B), and (C) the pulse width is the same, even though the carrier frequency is different. In these three cases average power would be the same. This is because average power is actually equal to the peak power of a pulse averaged over 1 operating cycle. However, the pulse width is increased in view (D) and we have a greater average power with the same prt. In view (E) the decreased pulse width has decreased average power over the same prt.
2-38
Use these simple rules to determine power in a pulsed-wave shape:
• Peak power is the maximum power reached by the transmitter during the pulse.
• Average power equals the peak power averaged over one cycle.
In pulse modulation you will need to know the percentage of time the system is actually producing rf. For example, let’s say that a pulse system is transmitting 25 percent of the time. This would mean that the pw is 1/4 the prt. For every 60 minutes we operate the pulse system, we actually transmit a total of only 15 minutes.
The DUTY CYCLE is the ratio of working time to total time for intermittently operated devices. Thus, duty cycle represents a ratio of actual transmitting time to transmitting time plus rest time. To establish the duty cycle, divide the pw by the prt of the system. This yields the duty cycle and is expressed as a decimal figure. With this information, we can figure percentage of transmitting time by multiplying the duty cycle by 100.
Pulse modulation has many applications in the transmission of intelligence information. In telemetry, for example, the width of successive pulses may tell us humidity; the changing of the rest time may tell us pressure. In other applications, as you will see later in this text, the changing of the average power can provide us with intelligence information.
In radar a pulse is transmitted and travels some distance to a target where it is then reflected back to the system. The amount of time it takes provides us with information that can be converted to distance.
Telemetry and radar systems use the principles of pulse modulation described in this section. Let’s quickly review what has been presented:
2-39
Pulse modulation will play a major part in your electronics career. In one way or another, you will encounter it in some form. The function of the particular system may involve many variations of the characteristics presented here. We will now look at some specific applications of pulse modulation in radar and communications systems.
Q-14. Overmodulating an rf carrier in amplitude modulation produces a waveform which is similar to what modulated waveform?
Q-15. What is prt? Q-16. What is nonpulse time? Q-17. What is average power in a pulsed system?
Radio frequency energy in radar is transmitted in short pulses with time durations that may vary from 1 to 50 microseconds or more. If the transmitter is cut off before any reflected energy returns from a target, the receiver can distinguish between the transmitted pulse and the reflected pulse. After all reflections have returned, the transmitter can again be cut on and the process repeated. The receiver output is applied to an indicator which measures the time interval between the transmission of energy and its return as a reflection. Since the energy travels at a constant velocity, the time interval becomes a measure of the distance traveled (RANGE). Since this method does not depend on the relative frequency of the returned signal, or on the motion of the target, difficulties experienced in cw or fm methods are not encountered. The pulse modulation method is used in many military radar applications.
Most radar oscillators operate at pulse voltages between 5 and 20 kilovolts. They require currents of several amperes during the actual pulse which places severe requirements on the modulator. The function of the high-vacuum tube modulator is to act as a switch to turn a pulse ON and OFF at the transmitter in response to a control signal. The best device for this purpose is one which requires the least signal power for control and allows the transfer of power from the transmitter power source to the oscillator with the least loss. The pulse modulator circuits discussed in this section are typical pulse modulators used in radar equipment.
The SPARK-GAP MODULATOR consists of a circuit for storing energy, a circuit for rapidly discharging the storage circuit (spark gap), a pulse transformer, and an ac power source. The circuit for storing energy is essentially a short section of artificial transmission line which is known as the PULSE-FORMING NETWORK (pfn). The pulse-forming network is discharged by a spark gap. Two types of spark gaps are used: FIXED GAPS and ROTARY GAPS. The fixed gap, discussed in this section, uses a trigger pulse to ionize the air between the contacts of the spark gap and to initiate the discharge of the pulse-forming network. The rotary gap is similar to a mechanically driven switch.
A typical fixed, spark-gap modulator circuit is shown in figure 2-38. Between trigger pulses the spark gap is an open circuit. Current flows through the pulse transformer (T1), the pulse-forming network (C1, C2, C3, C4, and L2), the diode (V1), and the inductor (L1) to the plate supply voltage (Eb). These components form the charging circuit for the pulse-forming network.
2-40
The spark gap is actually triggered (ionized) by the combined action of the charging voltage across the pulse-forming network and the trigger pulse. (Ionization was discussed in NEETS, Module 6, Introduction to Electronic Emission, Tubes and Power Supplies.) The air between the trigger pulse injection point and ground is ionized by the trigger voltage. This, in turn, initiates the ionization of the complete gap by the charging voltage. This ionization allows conduction from the charged pulse-forming network through pulse transformer T1. The output pulse is then applied to an oscillating device, such as a magnetron.
The hydrogen THYRATRON MODULATOR is an electronic switch which requires a positive trigger of only 150 volts. The trigger potential must rise at the rate of 100 volts per microsecond to cause the modulator to conduct. In contrast to spark gap devices, the hydrogen thyratron (figure 2-39) operates over a wide range of anode voltages and pulse-repetition rates. The grid has complete control over the initiation of cathode emission for a wide range of voltages. The anode is completely shielded from the cathode by the grid. Thus, effective grid action results in very smooth firing over a wide range of anode voltages and repetition frequencies. Unlike most other thyratrons, the positive grid-control characteristic ensures stable operation. In addition, deionization time is reduced by using the hydrogen-filled tube.
The hydrogen thyratron modulator provides improved timing because the synchronized trigger pulse is applied to the control grid of the thyratron (V2) and instantaneous firing is obtained. In addition, only
2-41
one gas tube is required to discharge the pulse-forming network, and a low amplitude trigger pulse is sufficient to initiate discharge. A damping diode is used to prevent breakdown of the thyratron by reverse-voltage transients. The thyratron requires a sharp leading edge for a trigger pulse and depends on a sudden drop in anode voltage (controlled by the pulse-forming network) to terminate the pulse and cut off the tube.
As shown in figure 2-39, the typical thyratron modulator is very similar to the spark-gap modulator. It consists of a power source (Eb), a circuit for storing energy (L2, C2, C3, C4, and C5), a circuit for discharging the storage circuit (V2), and a pulse transformer (T1). In addition this circuit has a damping diode (V1) to prevent reverse-polarity signals from being applied to the plate of V2 which could cause V2 to breakdown.
With no trigger pulse applied, the pfn charges through T1, the pfn, and the charging coil L1 to the potential of Eb. When a trigger pulse is applied to the grid of V2, the tube ionizes causing the pulse- forming network to discharge through V2 and the primary of T1. As the voltage across the pfn falls below the ionization point of V2, the tube shuts off. Because of the inductive properties of the pfn, the positive discharge voltage has a tendency to swing negative. This negative overshoot is prevented from damaging the thyratron and affecting the output of the circuit by V1, R1, R2, and C1. This is a damping circuit and provides a path for the overshoot transient through V1. It is dissipated by R1 and R2 with C1 acting as a high-frequency bypass to ground, preserving the sharp leading and trailing edges of the pulse. The hydrogen thyratron modulator is the most common radar modulator.
Pulse modulation is also useful in communications systems. The intelligence-carrying capability and power requirements for communications systems differ from those of radar. Therefore, other methods of achieving pulse modulation that are more suitable for communications systems will now be studied.
Q-18. What is the primary component for a spark-gap modulator? Q-19. What are the basic components of a thyratron modulator?
To transmit intelligence using pulse modulation, you must provide a method to vary some characteristic of the pulse train in accordance with the modulating signal. Figure 2-40 illustrates a simple pulse train. The characteristics of these pulses that can be varied are amplitude, pulse width, pulse-repetition time, and the pulse position as compared to a reference. In addition to these three characteristics, pulses may be transmitted according to a code to represent the different levels of the modulating signal. To ensure maximum fidelity (accuracy in reproducing a modulating wave), the modulating signal has to be represented by enough pulses to restore the original wave shape. Logically, the higher the sampling rate (the more often sampled) of the pulse modulator, the more accurately the original modulating wave can be reproduced. Figure 2-41 illustrates the effectiveness of three pulse-sampling rates. View (A) shows a sampling rate of more than two times the modulating frequency. As you can see, this reproduces the modulating signal very accurately. However, the high sampling rate requires a wide bandwidth and increases the average power required of the transmitter. If less than two samples per cycle are made, you are not able to reproduce the original modulating signal, as shown in view (B). View (C) shows a sampling rate that is two times the highest modulating frequency.  The standard sampling rate is 2.5 times the highest frequency that is to be transmitted. This ensures the ability to accurately reproduce the modulating waveform. In military voice systems the bandwidth for voice signals is limited to 300 to 3,000 hertz, requiring a sampling frequency of 8 kilohertz. Although the pulse characteristic that is changed may vary for each type of pulse modulation, the sampling frequency will remain constant. We will now briefly discuss common types of pulse modulation.
2-43
Some characteristic of the sampling pulses must be varied by the modulating signal for the intelligence of the signal to be present in the pulsed wave. Figure 2-42 shows three typical waveforms in which the pulse amplitude is varied by the amplitude of the modulating signal. View (A) represents a sine wave of intelligence to be modulated on a transmitted carrier wave. View (B) shows the timing pulses which determine the sampling interval. View (C) shows PULSE-AMPLITUDE MODULATION (pam) in which the amplitude of each pulse is controlled by the instantaneous amplitude of the modulating signal at the time of each pulse.
Pulse-amplitude modulation is the simplest form of pulse modulation. It is generated in much the same manner as analog-amplitude modulation. The timing pulses are applied to a pulse amplifier in which the gain is controlled by the modulating waveform. Since these variations in amplitude actually represent the signal, this type of modulation is basically a form of AM. The only difference is that the signal is now in the form of pulses. This means that pam has the same built-in weaknesses as any other AM signal - high susceptibility to noise and interference. The reason for susceptibility to noise is that any interference in the transmission path will either add to or subtract from any voltage already in the circuit (signal voltage). Thus, the amplitude of the signal will be changed. Since the amplitude of the voltage represents the signal, any unwanted change to the signal is considered a SIGNAL DISTORTION. For this reason, pam is not often used. When pam is used, the pulse train is used to frequency modulate a carrier for transmission. Techniques of pulse modulation other than pam have been developed to overcome problems of noise interference. The following sections will discuss other types of pulse modulation.
Q-20. What action is necessary to impress intelligence on the pulse train in pulse modulation?
2-44
Q-21. To ensure the accuracy of a transmission, what is the minimum number of times a modulating wave should be sampled in pulse modulation?
Q-22. What, if any, noise susceptibility advantage exists for pulse-amplitude modulation over analog- amplitude modulation?
In pulse-modulated systems, as in an analog system, the intelligence may be impressed on the carrier by varying any of its characteristics. In the preceding paragraphs the method of modulating a pulse train by varying its amplitude was discussed. Time characteristics of pulses may also be modulated with intelligence information. Two time characteristics may be affected: (1) the time duration of the pulses, referred to as PULSE-DURATION MODULATION (pdm) or PULSE-WIDTH MODULATION (pwm); and (2) the time of occurrence of the pulses, referred to as PULSE-POSITION MODULATION (ppm), and a special type of PULSE-TIME MODULATION (ptm) referred to as PULSE-FREQUENCY MODULATION (pfm). Figure 2-43 shows these types of ptm in views (C), (D), and (E). Views (A) and (B) show the modulating signal and timing, respectively.
2-45
PULSE-DURATION MODULATION.—Pdm and pwm are designations for a single type of modulation. The width of each pulse in a train is made proportional to the instantaneous value of the modulating signal at the instant of the pulse. Either the leading edges, the trailing edges, or both edges of the pulses may be modulated to produce the variation in pulse width. Pdm can be obtained in a number of ways, one of which is illustrated in views (A) through (D) in figure 2-44. A circuit to produce pdm is shown in figure 2-45. Adding the modulating signal [figure 2-44, view (A)] to a repetitive sawtooth [view (B)] will result in the waveform shown in view (C). This waveform is then applied to a circuit which changes state when the input signal exceeds a specific threshold level. This action produces pulses with widths that are determined by the length of time that the input waveform exceeds the threshold level. The resulting waveform is shown in view (D).
2-46
In the circuit of figure 2-45, a series of sawtooth pulses, occurring at the sampling rate, is applied to a one-shot multivibrator. The multivibrator has the signal voltage Essuperimposed on the bias voltage Ein. Each pulse triggers a cycle of multivibrator operation which terminates after a time interval and varies linearly with the voltage Es. The pulse of plate voltage produced by the multivibrator will have a leading edge at T1. The leading edge will vary in position with the signal voltage, while the trailing edge at T2 is fixed by the termination of the sawtooth pulse. The length of the output pulse is thus duration or width modulated. If the sawtooth has an instantaneous buildup and a sloping trailing edge, then the leading edge (T1) is fixed and the trailing edge (T2) varies. If the sawtooth generator produces a slope on both leading and trailing edges, both T1 and T2 are variable in position, but the result is still pdm. Pdm is often used because it is of a constant amplitude and is, therefore, less susceptible to noise. When compared with ppm, pdm has the disadvantage of a varying pulse, width and, therefore, of varying power content. This means that the transmitter must be powerful enough to handle the maximum-width pulses, although the average power transmitted is much less than peak power. On the other hand, pdm will still work if the synchronization between the transmitter and receiver fails; in ppm it will not, as will be seen in the next section.
PULSE-POSITION MODULATION.—The amplitude and width of the pulse is kept constant in the system. The position of each pulse, in relation to the position of a recurrent reference pulse, is varied by each instantaneous sampled value of the modulating wave. Ppm has the advantage of requiring constant transmitter power since the pulses are of constant amplitude and duration. It is widely used but has the disadvantage of depending on transmitter-receiver synchronization.
2-47
Ppm can be generated in several ways, but we will discuss one of the simplest. Figure 2-46 shows three waveforms associated with developing ppm from pdm. The pdm pulse train is applied to a differentiating circuit. (Differentiation was presented in NEETS, Module 9, Introduction to Wave-Generation and Wave-Shaping Circuits.) This provides positive- and negative-polarity pulses that correspond to the leading and trailing edges of the pdm pulses. Considering pdm and its generation, you can see that each pulse has a leading and trailing edge. In this case the position of the leading edge is fixed, whereas the trailing edge is not, as shown in view (A) of figure 2-46. The resultant pulses after the differentiation are shown in view (B). The negative pulses are position-modulated in accordance with the modulating waveform. Both the negative and positive pulse are then applied to a rectification circuit. This application eliminates the positive, non-modulated pulses and develops a ppm pulse train, as shown in view (C).
PULSE-FREQUENCY MODULATION.—Pfm is a method of pulse modulation in which the modulating wave is used to frequency modulate a pulse-generating circuit. For example, the pulse rate may be 8,000 pulses per second (pps) when the signal voltage is 0. The pulse rate may step up to 9,000 pps for maximum positive signal voltage, and down to 7,000 pps for maximum negative signal voltage. Figure 2-47, views (A), (B), and (C) show three typical waveforms for pfm. This method of modulation is not used extensively because of complicated pfm generation methods. It requires a stable oscillator that is frequency modulated to drive a pulse generator. Since the other forms of ptm are easier to achieve, they are commonly used.
2-48
Q-23. What characteristics of a pulse can be changed in pulse-time modulation? Q-24. Which edges of the pulse can be modulated in pulse-duration modulation? Q-25. What is the main disadvantage of pulse-position modulation? Q-26. What is pulse-frequency modulation?
PULSE-CODE MODULATION (pcm) refers to a system in which the standard values of a QUANTIZED WAVE (explained in the following paragraphs) are indicated by a series of coded pulses. When these pulses are decoded, they indicate the standard values of the original quantized wave. These codes may be binary, in which the symbol for each quantized element will consist of pulses and spaces: ternary, where the code for each element consists of any one of three distinct kinds of values (such as positive pulses, negative pulses, and spaces); or n-ary, in which the code for each element consists of nay number (n) of distinct values. This discussion will be based on the binary pcm system.
All of the pulse-modulation systems discussed previously provide methods of converting analog wave shapes to digital wave shapes (pulses occurring at discrete intervals, some characteristic of which is varied as a continuous function of the analog wave). The entire range of amplitude (frequency or phase) values of the analog wave can be arbitrarily divided into a series of standard values. Each pulse of a pulse train [figure 2-48, view (B)] takes the standard value nearest its actual value when modulated. The modulating wave can be faithfully reproduced, as shown in views (C) and (D). The amplitude range has been divided into 5 standard values in view (C). Each pulse is given whatever standard value is nearest its actual instantaneous value. In view (D), the same amplitude range has been divided into 10 standard levels. The curve of view (D) is a much closer approximation of the modulating wave, view (A), than is the 5-level quantized curve in view (C). From this you should see that the greater the number of standard levels used, the more closely the quantized wave approximates the original. This is also made evident by the fact that an infinite number of standard levels exactly duplicates the conditions of nonquantization (the original analog waveform).
2-49
Although the quantization curves of figure 2-48 are based on 5- and 10-level quantization, in actual practice the levels are usually established at some exponential value of 2, such as 4(22), 8(2 3), 16(24), 32(25) . . . N(2n). The reason for selecting levels at exponential values of 2 will become evident in the discussion of pcm. Quantized fm is similar in every way to quantized AM. That is, the range of frequency deviation is divided into a finite number of standard values of deviation. Each sampling pulse results in a deviation equal to the standard value nearest the actual deviation at the sampling instant. Similarly, for phase modulation, quantization establishes a set of standard values. Quantization is used mostly in amplitude- and frequency-modulated pulse systems.
Figure 2-49 shows the relationship between decimal numbers, binary numbers, and a pulse-code waveform that represents the numbers. The table is for a 16-level code; that is, 16 standard values of a quantized wave could be represented by these pulse groups. Only the presence or absence of the pulses are important. The next step up would be a 32-level code, with each decimal number represented by a series of five binary digits, rather than the four digits of figure 2-49. Six-digit groups would provide a 64-level code, seven digits a 128-level code, and so forth. Figure 2-50 shows the application of pulse-coded groups to the standard values of a quantized wave.
2-50
In figure 2-50 the solid curve represents the unquantized values of a modulating sinusoid. The dashed curve is reconstructed from the quantized values taken at the sampling interval and shows a very close agreement with the original curve. Figure 2-51 is identical to figure 2-50 except that the sampling interval is four times as great and the reconstructed curve is not faithful to the original. As previously stated, the sampling rate of a pulsed system must be at least twice the highest modulating frequency to get
2-51
a usable reconstructed modulation curve. At the sampling rate of figure 2-50 and with a 4-element binary code, 128 bits (presence or absence of pulses) must be transmitted for each cycle of the modulating frequency. At the sampling rate of figure 2-51, only 32 bits are required; at the minimum sampling rate, only 8 bits are required.
As a matter of convenience, especially to simplify the demodulation of pcm, the pulse trains actually transmitted are reversed from those shown in figures 2-49, 2-50, and 2-51; that is, the pulse with the lowest binary value (least significant digit) is transmitted first and the succeeding pulses have increasing binary values up to the code limit (most significant digit). Pulse coding can be performed in a number of ways using conventional circuitry or by means of special cathode ray coding tubes. One form of coding circuit is shown in figure 2-52. In this case, the pulse samples are applied to a holding circuit (a capacitor which stores pulse amplitude information) and the modulator converts pam to pdm. The pdm pulses are then used to gate the output of a precision pulse generator that controls the number of pulses applied to a binary counter. The duration of the gate pulse is not necessarily an integral number of the repetition pulses from the precisely timed clock-pulse generator. Therefore, the clock pulses gated into the binary counter by the pdm pulse may be a number of pulses plus the leading edge of an additional pulse. This "partial" pulse may have sufficient duration to trigger the counter, or it may not. The counter thus responds only to integral numbers, effectively quantizing the signal while, at the same time, encoding it. Each bistable stage of the counter stores ZERO or a ONE for each binary digit it represents (binary 1110 or decimal 14 is shown in figure 2-52). An electronic commutator samples the 20, 21, 22, and 23 digit positions in sequence and transmits a mark or space bit (pulse or no pulse) in accordance with the state of each counter stage. The holding circuit is always discharged and reset to zero before initiation of the sequence for the next pulse sample.
2-52
The pcm demodulator will reproduce the correct standard amplitude represented by the pulse-code group. However, it will reproduce the correct standard only if it is able to recognize correctly the presence or absence of pulses in each position. For this reason, noise introduces no error at all if the signal-to-noise ration is such that the largest peaks of noise are not mistaken for pulses. When the noise is random (circuit and tube noise), the probability of the appearance of a noise peak comparable in amplitude to the pulses can be determined. This probability can be determined mathematically for any ration of signal-to-average- noise power. When this is done for 10 5 pulses per second, the approximate error rate for three values of signal power to average noise power is:
Above a threshold of signal-to-noise ration of approximately 20 dB, virtually no errors occur. In all other systems of modulation, even with signal-to-noise ratios as high as 60 dB, the noise will have some effect. Moreover, the pcm signal can be retransmitted, as in a multiple relay link system, as many times as desired, without the introduction of additional noise effects; that is, noise is not cumulative at relay stations as it is with other modulation systems.
The system does, of course, have some distortion introduced by quantizing the signal. Both the standard values selected and the sampling interval tend to make the reconstructed wave depart from the original. This distortion, called QUANTIZING NOISE, is initially introduced at the quantizing and coding modulator and remains fixed throughout the transmission and retransmission processes. Its magnitude can be reduced by making the standard quantizing levels closer together. The relationship of the quantizing noise to the number of digits in the binary code is given by the following standard relationship:
Where:
2-53
Thus, with the 4-digit code of figure 2-50 and 2-51, the quantizing noise will be about 35 dB weaker than the peak signal which the channel will accommodate.
The advantages of pcm are two-fold. First, noise interference is almost completely eliminated when the pulse signals exceed noise levels by a value of 20 dB or more. Second, the signal may be received and retransmitted as many times as may be desired without introducing distortion into the signal.
Q-27. Pulse-code modulation requires the use of approximations of value that are obtained by what process?
Q-28. If a modulating wave is sampled 10 times per cycle with a 5-element binary code, how many bits of information are required to transmit the signal?
Q-29. What is the primary advantage of pulse-modulation systems?
Now that you have completed this chapter, a short review of what you have learned is in order. The following review will refresh your memory of angle modulation and pulse modulation.
FREQUENCY-SHIFT KEYING (fsk) is similar to cw keying in amplitude modulation and is a form of angle modulation. The carrier frequency is changed between two discrete values by the opening and closing of a key.
2-54
In FREQUENCY MODULATION (fm) the instantaneous frequency of the radio-frequency wave is varied in accordance with the modulating signal; the amplitude of the radio-frequency wave is kept constant.
2-55
MODULATION INDEX is the ratio of the maximum frequency difference between the modulated and the unmodulated carrier, or between the deviation frequency and the modulation frequency.
The number of SIGNIFICANT SIDEBANDS and the modulating frequency will determine the bandwidth of the fm wave. The number of significant sidebands can be determined from the modulation index.
.01 2 .4 2 .5 4
1.0 6 2.0 8 3.0 12 4.0 14 5.0 16 6.0 18 7.0 22 8.0 24 9.0 26
10.0 28 11.0 32 12.0 32 13.0 36 14.0 38 15.0 38
The REACTANCE-TUBE MODULATOR is frequency modulated by using a reactance tube in shunt with the tank circuit.
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The SEMICONDUCTOR-REACTANCE MODULATOR is used to frequency modulate low-power semiconductor transmitters.
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The MULTIVIBRATOR MODULATOR uses an astable multivibrator with a modulating voltage inserted in series with the base return of the multivibrator transistors.
The VARACTOR FM MODULATOR uses a VARACTOR. This is a specially designed diode that has a certain amount of capacitance between the junction that can be controlled by reverse biasing.
In PHASE MODULATION the carrier’s phase is caused to shift at the rate of the modulating audio. The amount of phase shift is controlled by the amplitude of the modulating wave.
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A BASIC PHASE MODULATOR may be a single tube in series with a capacitor to form a phase-shift network. As the impedance of the tube changes, the phase of the output shifts.
PHASE-SHIFT KEYING (psk) is similar to cw and fsk. It consists of phase reversals of the carrier frequency as modulating signal data elements open and close the modulator key.
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PULSE MODULATION is modulation in which we allow oscillations to occur for a given period of time only during selected intervals.
PULSE-REPETITION TIME (prt) is the specific time period between each group of rf pulses.
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PULSE-REPETITION FREQUENCY (prf) is found by dividing the pulse repetition time into 1. This defines how often the groups of pulses occur.
PULSE WIDTH (pw) or PULSE DURATION (pd) is the time that a pulse is occurring.
REST TIME (rt) is the time referred to as nonpulse time.
PEAK POWER is the maximum power during a pulse.
AVERAGE POWER equals the peak power averaged over one complete cycle.
DUTY CYCLE is the ratio of working time to total time, or the ratio of actual transmit time to transmit time plus rest time, for intermittently operated devices.
The SPARK-GAP MODULATOR consists of a circuit for storing energy, a circuit for rapidly discharging the storage circuit, a pulse transformer, and a power source.
The THYRATRON MODULATOR is an electronic switch which requires a positive trigger of only 150 volts. The trigger must rise at the rate of 100 volts per microsecond to fire or cause the modulator to conduct.
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In communications PULSE-MODULATION SYSTEMS, the modulating wave must be SAMPLED at 2.5 times the highest modulating frequency to ensure accuracy.
PULSE-AMPLITUDE MODULATION (pam) is modulation in which the amplitude of each pulse is controlled by the instantaneous amplitude of the modulation signal at the time of each pulse.
PULSE-DURATION MODULATION (pdm) or PULSE-WIDTH MODULATION (pwm) are both designations for a type of modulation. The width of each pulse in a train is made proportional to the instantaneous value of the modulating signal at the instant of the pulse.
PULSE-POSITION MODULATION (ppm) has the advantage of requiring constant transmitter power. The amplitude and width of the pulses are kept constant. At the same time, the position of each pulse, in relation to the position of a recurrent reference pulse, is varied by each instantaneous sampled value of the modulating wave.
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PULSE-FREQUENCY MODULATION (pfm) is a method of pulse modulation in which the modulating wave is used to frequency modulate a pulse-generating circuit.
PULSE-CODE MODULATION (pcm) refers to a system in which the standard value of a quantized wave is indicated by a series of coded pulses that give the modulating wave’s value at the instant of the sample.
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ANSWERS TO QUESTIONS Q1. THROUGH Q29. A-1. Frequency and phase.
A-2. Frequency-shift keying. A-3. Resistance to noise interference. A-4. Instantaneous frequency. A-5. As the ratio of the frequency deviation to the maximum frequency deviation allowable. A-6. The number of significant sidebands and the modulating frequency. A-7. By changing the reactance of an oscillator circuit in consonance with the modulating voltage. A-8. Collector-to-emitter capacitance.
A-9. An LCR filter. A-10. Capacitance.
A-11. Phase.
A-12. A phase-shift network such as a variable resistor and capacitor in series. A-13. Cw and frequency-shift keying. A-14. Pulse modulation.
A-15. Pulse-repetition time.
A-16. Rest time.
A-17. Peak power during a pulse averaged over pulse time plus rest time.
A-18. Either a fixed spark gap that uses a trigger pulse to ionize the air between the contacts, or a rotary gap that is similar to a mechanical switch.
A-19. Power source, a circuit for storing energy, a circuit for discharging the storage circuit, and a pulse transformer.
A-20. Some characteristic of the pulses has to be varied. A-21. 2.5 times the highest modulating frequency. A-22. Both are susceptible to noise and interference. A-23. The time duration of the pulses or the time of occurrence of the pulses. A-24. Either, or both at the same time.
A-25. It requires synchronization between the transmitter and receiver.
A-26. A method of pulse modulation in which a modulating wave is used to frequency modulate a pulse-generating circuit.
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Upon completion of this chapter you will be able to:
1. Describe cw detector circuit operations for the heterodyne and regenerative detectors.
2. Discuss the requirements for recovery of intelligence from an AM signal and describe the theory of operation of the following AM demodulators: series-diode, shunt-diode, common-emitter, and common-base.
3. Describe fm demodulation circuit operation for the phase-shift and gated-beam discriminators and the ratio-detector demodulator.
4. Describe phase demodulation circuit operation for the peak, low-pass filter, and conversion detectors.
In chapters 1 and 2 you studied how to apply intelligence (modulation) to an rf-carrier wave. Carrier modulation allows the transmission of modulating frequencies without the use of transmission wire as a medium. However, for the communication process to be completed or to be useful, the intelligence must be recovered in its original form at the receiving site. The process of re-creating original modulating frequencies (intelligence) from the rf carrier is referred to as DEMODULATION or DETECTION. Each type of modulation is different and requires different techniques to recover (demodulate) the intelligence. In this chapter we will discuss ways of demodulating AM, cw, fm, phase, and pulse modulation.
The circuit in which restoration is achieved is called the DETECTOR or DEMODULATOR (both of these terms are used in NEETS). The term demodulator is used because the demodulation process is considered to be the opposite of modulation. The output of an ideal detector must be an exact reproduction of the modulation existing on the rf wave. Failure to accurately recover this intelligence will result in distortion and degradation of the demodulated signal and intelligence will be lost. The distortion may be in amplitude, frequency, or phase, depending on the nature of the demodulator. A nonlinear device is required for demodulation. This nonlinear device is required to recover the modulating frequencies from the rf envelope. Solid-state detector circuits may be either a pn junction diode or the input junction of a transistor. In electron-tube circuits, either a diode or the grid or plate circuits of a triode electron tube may be used as the nonlinear device.
Q-1. What is demodulation? Q-2. What is a demodulator?
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Continuous-wave (cw) modulation consists of on-off keying of a carrier wave. To recover on-off keyed information, we need a method of detecting the presence or absence of rf oscillations. The CW DEMODULATOR detects the presence of rf oscillations and converts them into a recognizable form. Figure 3-1 illustrates the received cw in view (A), the rectified cw from a diode detector in view (B), and the dc output from a filter that can be used to control a relay or light indicator in view (C).
Figure 3-2 is a, simplified circuit that could be used as a cw demodulator. The antenna receives the rf oscillations from the transmitter. The tank circuit, L and C1, acts as a frequency-selective network that is tuned to the desired rf carrier frequency. The diode rectifies the oscillations and C2 provides filtering to provide a constant dc output to control the headset. This demodulator circuit is the equivalent of a wire telegraphy circuit but it has certain disadvantages. For example, if two transmitters are very close in frequency, distinguishing which transmitting station you are receiving is often impossible without a method of fine tuning the desired frequency. Also, if the stations are within the frequency bandpass of the input tank circuit, the tank output will contain a mixture of both signals. Therefore, a method, such as HETERODYNE DETECTION, must be used which provides more than just the information on the presence or absence of a signal.
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The use of an af voltage in the detector aids the operator in distinguishing between various signals. Since the carrier is unmodulated, the af voltage can be developed by using the heterodyne procedure discussed in chapter 1. The procedure is to mix the incoming cw signal with locally generated oscillations. This provides a convenient difference frequency in the af range, such as 1,000 hertz. The af difference frequency then is rectified and smoothed by a detector. The af voltage is reproduced by a telephone headset or a loudspeaker.
Consider the heterodyne reception of the code letter A, as shown in figure 3-3, view (A). The code consists of a short burst of cw energy (dot) followed by a longer burst (dash). Assume that the frequency of the received cw signal is 500 kilohertz. The locally generated oscillations are adjusted to a frequency which is higher or lower than the incoming rf signal (501 kilohertz in this case), as shown in view (B). The voltage resulting from the heterodyning action between the cw signal [view (A)] and the local oscillator signal [view (B)] is shown in view (C) as the mixed-frequency signal. ENVELOPE (intelligence) amplitude varies at the BEAT (difference) frequency of 1,000 hertz (501,000 - 500,000). The negative half cycles of the mixed frequency are rectified, as shown in view (D). The peaks of the positive half cycles follow the 1,000-hertz beat frequency.
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The cw signal pulsations are removed by the rf filter in the detector output and only the envelope of the rectified pulses remains. The envelope, shown in view (E), is a 1,000-hertz audio-beat note. This 1,000 hertz, dot-dash tone may be heard in a speaker or headphone and identified as the letter A by the operator.
The heterodyne method of reception is highly selective and allows little interference from adjacent cw stations. If a cw signal from a radiotelegraph station is operating at 10,000,000 hertz and at the same time an adjacent station is operating at 10,000,300 hertz, a simple detector cannot clearly discriminate between the two stations because the signals are just 300 hertz apart. This is because the bandpass of the tuning circuits is too wide and allows some of the other signal to interfere. The two carrier frequencies differ by only 0.003 percent and a tuned tank circuit cannot easily discriminate between them. However, if a heterodyne detector with a local-oscillator frequency of 10,001,000 hertz is used, then beat notes of 1,000 and 700 hertz are produced by the two signals. These are audio frequencies, which can be
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distinguished easily by a selective circuit because they differ by 30 percent (compared to the 0.003 percent above).
Even if two stations produce identical beat frequencies, they can be separated by adjusting the local-oscillator or BEAT-FREQUENCY OSCILLATOR (bfo) frequency. For example, if the second station in the previous example had been operating at 10,002,000 hertz, then both stations would have produced a 1,000-hertz beat frequency and interference would have occurred. Adjusting the local-oscillator frequency to 9,999,000 hertz would have caused the desired station at 10,000,000 hertz to produce a 1,000-hertz beat frequency. The other station, at 10,002,000 hertz, would have produced a beat frequency of 3,000 hertz. Either selective circuits or the operator can easily distinguish between these widely differing tones. A trained operator can use the variable local oscillator to distinguish between stations that vary in frequency by only a few hundred hertz.
Q-3. What is the simplest form of cw detector? Q-4. What are the essential components of a cw receiver system? Q-5. What principle is used to help distinguish between two cw signals that are close in frequency? Q-6. How does heterodyning distinguish between cw signals?
A simple, one-transistor REGENERATIVE DETECTOR circuit that uses the heterodyning principle for cw operation is shown in figure 3-4. The circuit can be made to oscillate by increasing the amount of energy fed back to the tank circuit from the collector-output circuit (by physically moving tickler coil L2 closer to L1 using the regeneration control). This feedback overcomes losses in the base-input circuit and causes self-oscillations which are controlled by tuning capacitor C1. The received signal from the antenna and the oscillating frequency are both present at the base of transistor Q1. These two frequencies are heterodyned by the nonlinearity of the transistor. The resulting beat frequencies are then rectified by the emitter-base junction and produce a beat note which is amplified in the collector-output circuit. The af currents in the collector circuit actuate the phones. The REGENERATIVE DETECTOR (figure 3-4) produces its own oscillations, heterodynes them with an incoming signal, and rectifies or detects them.
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The regenerative detector is used to receive short-wave code signals because it is easy to adjust and has high sensitivity and good selectivity. At high frequencies, the amount of signal detuning necessary to produce an audio-beat note is a small percentage of the signal frequency and causes no trouble. The use of the regenerative detector for low-frequency code reception, however, is usually avoided. At low frequencies the detuning required to produce the proper audio-beat frequency is a considerable percentage of the signal frequency. Although this type detector may be used for AM signals, it has high distortion and is not often used.
Q-7. What simple, one-transistor detector circuit uses the heterodyne principle? Q-8. What three functions does the transistor in a regenerative detector serve?
Amplitude modulation refers to any method of modulating an electromagnetic carrier frequency by varying its amplitude in accordance with the message intelligence that is to be transmitted. This is accomplished by heterodyning the intelligence frequency with the carrier frequency. The vector summation of the carrier, sum, and difference frequencies causes the modulation envelope to vary in amplitude at the intelligence frequency, as discussed in chapter 1. In this section we will discuss several circuits that can be used to recover this intelligence from the variations in the modulation envelope.
The detection of AM signals ordinarily is accomplished by means of a diode rectifier, which may be either a vacuum tube or a semiconductor diode. The basic detector circuit is shown in its simplest form in view (A) of figure 3-5. Views (B), (C), and (D) show the circuit waveforms. The demodulator must meet three requirements: (1) It must be sensitive to the type of modulation applied at the input, (2) it must be nonlinear, and (3) it must provide filtering. Remember that the AM waveform appears like the diagram of view (B) and the amplitude variations of the peaks represent the original audio signal, but no modulating signal frequencies exist in this waveform. The waveform contains only three rf frequencies: (1) the carrier frequency, (2) the sum frequency, and (3) the difference frequency. The modulating intelligence is contained in the difference between these frequencies. The vector addition of these frequencies provides the modulation envelope which approximates the original modulating waveform. It is this modulation envelope that the DIODE DETECTORS use to reproduce the original modulating frequencies.
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Let’s analyze the operation of the circuit shown in view (A) of figure 3-5. This circuit is the basic type of diode receiver and is known as a SERIES-DIODE DETECTOR. The circuit consists of an antenna, a tuned LC tank circuit, a semiconductor diode detector, and a headset which is bypassed by capacitor C2. The antenna receives the transmitted rf energy and feeds it to the tuned tank circuit. This tank circuit (L1 and C1) selects which rf signal will be detected. As the tank resonates at the selected frequency, the wave shape in view (B) is developed across the tank circuit. Because the semiconductor is a nonlinear device, it conducts in only one direction. This eliminates the negative portion of the rf carrier and produces the signal shown in view (C). The current in the circuit must be smoothed before the headphones can reproduce the af intelligence. This action is achieved by C2 which acts as a filter to
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provide an output that is proportional to the peak rf pulses. The filter offers a low impedance to rf and a relatively high impedance to af. (Filters were discussed in NEETS, Module 9, Introduction to Wave- Generation and Wave-Shaping Circuits.) This action causes C2 to develop the waveform in view (D). This varying af voltage is applied to the headset which then reproduces the original modulating frequency. This circuit is called a series-diode detector (sometimes referred to as a VOLTAGE-DIODE DETECTOR) because the semiconductor diode is in series with both the input voltage and the load impedance. Voltages in the circuit cause an output voltage to develop across the load impedance that is proportional to the input voltage peaks of the modulation envelope.
Q-9. What are the three requirements for an AM demodulator? Q-10. What does the simplest diode detector use to reproduce the modulating frequency? Q-11. What is the function of the diode in a series-diode detector? Q-12. In figure 3-5, what is the function of C2?
The SHUNT-DIODE DETECTOR (figure 3-6) is similar to the series-diode detector except that the output variations are current pulses rather than voltage pulses. Passing this current through a shunt resistor develops the voltage output. The input is an rf modulated envelope. On the negative half cycles of the rf, diode CR1 is forward biased and shunts the signal to ground. On the positive half cycles, current flows from the output through L1 to the input. A field is built up around L1 that tends to keep the current flowing. This action integrates the rf current pulses and causes the output to follow the modulation envelope (intelligence) closely. (Integration was discussed in NEETS, Module 9, Introduction to Wave-Generation and Wave-Shaping Circuits.) Shunt resistor R1 develops the output voltage from this current flow. Although the shunt detector operates on the principle of current flow, it is the output voltage across the shunt resistor that is used to reproduce the original modulation signal. The shunt-diode detector is easily identified by noting that the detector diode is in parallel with both the input and load impedance. The waveforms associated with this detector are identical to those shown in views (B), (C), and (D) of figure 3-5.
The series-diode detector is normally used where large input signals are supplied and a linear output is required. The shunt-diode detector is used where the voltage variations are too small to produce a full output from audio amplifier stages. Additional current amplifiers are required to bring the output to a usable level. Other methods of detection and amplification have been developed which will detect low-
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level signals. The next sections will discuss two of these circuits, the common-emitter and common-base detectors.
Q-13. How does the current-diode detector differ from the voltage-diode detector? Q-14. Under what circuit conditions would the shunt detector be used?
The COMMON-EMITTER DETECTOR is often used in receivers to supply an amplified detected output. The schematic for a typical transistor common-emitter detector is shown in figure 3-7. Input transformer T1 has a tuned primary that acts as a frequency-selective device. L2 inductively couples the input modulation envelope to the base of transistor Q1. Resistors R1 and R2 are fixed-bias voltage dividers that set the bias levels for Q1. Resistor R1 is bypassed by C2 to eliminate rf. This RC combination also acts as the load for the diode detector (emitter-base junction of Q1). The detected audio is in series with the biasing voltage and controls collector current. The output is developed across R4 which is also bypassed to remove rf by C4. R3 is a temperature stabilization resistor and C3 bypasses it for both rf and af.
Q1 is biased for slight conduction with no input signal applied. When an input signal appears on the base of Q1, it is rectified by the emitter-base junction (operating as a diode) and is developed across R1 as a dc bias voltage with a varying af component. This voltage controls bias and collector current for Q1. The output is developed by collector current flow through R4. Any rf ripple in the output is bypassed across the collector load resistor by capacitor C4. The af variations are not bypassed. After the modulation envelope is detected in the base circuit, it is amplified in the output circuit to provide suitable af output. The output of this circuit is higher than is possible with a simple detector. Because of the amplification in this circuit, weaker signals can be detected than with a simple detector. A higher, more usable output is thus developed.
Q-15. Which junction of the transistor in the common-emitter detector detects the modulation envelope?
Q-16. Which component in figure 3-7 develops the af signal at the input? Q-17. How is the output signal developed in the common-emitter detector?
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Another amplifying detector that is used in portable receivers is the COMMON-BASE DETECTOR. In this circuit detection occurs in the emitter-base junction and amplification occurs at the output of the collector junction. The output developed is the equivalent of a diode detector which is followed by a stage of audio amplification, but with more distortion. Figure 3-8 is a schematic of a typical common-base detector. Transformer T1 is tuned by capacitor C3 to the frequency of the incoming modulated envelope. Resistor R1 and capacitor C1 form a self-biasing network which sets the dc operating point of the emitter junction. The af output is taken from the collector circuit through audio transformer T2. The primary of T2 forms the detector output load and is bypassed for rf by capacitor C2.
The input signal is coupled through T1. When capacitor C3 is tuned to the proper frequency, the signal is passed to the emitter of Q1. When no input signal is present, bias is determined by resistor R1. When the input signal becomes positive, current flows through the emitter-base junction causing it to be forward biased. C1 and R1 establish the dc operating point by acting as a filter network. This action provides a varying dc voltage that follows the peaks of the rf modulated envelope. This action is identical to the diode detector with the emitter-base junction doing the detecting. The varying dc voltage on the emitter changes the bias on Q1 and causes collector current to vary in accordance with the detected voltage. Transformer T2 couples these af current changes to the output. Thus, Q1 detects the AM wave and then provides amplification for the detected waveform.
The four AM detectors just discussed are not the only types that you will encounter. However, they are representative of most AM detectors and the same characteristics will be found in all AM detectors. Now let’s study some ways of demodulating frequency-modulated (fm) signals.
Q-18. Which junction acts as the detector in a common-base detector? Q-19. To what circuit arrangement is a common-base detector equivalent? Q-20. In figure 3-8, which components act as the filter network in the diode detector?
In fm demodulators, the intelligence to be recovered is not in amplitude variations; it is in the variation of the instantaneous frequency of the carrier, either above or below the center frequency. The
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detecting device must be constructed so that its output amplitude will vary linearly according to the instantaneous frequency of the incoming signal.
Several types of fm detectors have been developed and are in use, but in this section you will study three of the most common: (1) the phase-shift detector, (2) the ratio detector, and (3) the gated-beam detector.
To be able to understand the principles of operation for fm detectors, you need to first study the simplest form of frequency-modulation detector, the SLOPE DETECTOR. The slope detector is essentially a tank circuit which is tuned to a frequency either slightly above or below the fm carrier frequency. View (A) of figure 3-9 is a plot of voltage versus frequency for a tank circuit. The resonant frequency of the tank is the frequency at point 4. Components are selected so that the resonant frequency is higher than the frequency of the fm carrier signal at point 2. The entire frequency deviation for the fm signal falls on the lower slope of the bandpass curve between points 1 and 3. As the fm signal is applied to the tank circuit in view (B), the output amplitude of the signal varies as its frequency swings closer to, or further from, the resonant frequency of the tank. Frequency variations will still be present in this waveform, but it will also develop amplitude variations, as shown in view (B). This is because of the response of the tank circuit as it varies with the input frequency. This signal is then applied to the diode detector in view (C) and the detected waveform is the output. This circuit has the major disadvantage that any amplitude variations in the rf waveform will pass through the tank circuit and be detected. This disadvantage can be eliminated by placing a limiter circuit before the tank input. (Limiter circuits were discussed in NEETS, Module 9, Introduction to Wave-Generation and Wave-Shaping Circuits.) This circuit is basically the same as an AM detector with the tank tuned to a higher or lower frequency than the received carrier.
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Q-21. What is the simplest form of fm detector? Q-22. What is the function of an fm detector?
The FOSTER-SEELEY DISCRIMINATOR is also known as the PHASE-SHIFT DISCRIMINATOR. It uses a double-tuned rf transformer to convert frequency variations in the received fm signal to amplitude variations. These amplitude variations are then rectified and filtered to provide a dc output voltage. This voltage varies in both amplitude and polarity as the input signal varies in frequency. A typical discriminator response curve is shown in figure 3-10. The output voltage is 0 when the input frequency is equal to the carrier frequency (fr). When the input frequency rises above the center frequency, the output increases in the positive direction. When the input frequency drops below the center frequency, the output increases in the negative direction.
The output of the Foster-Seeley discriminator is affected not only by the input frequency, but also to a certain extent by the input amplitude. Therefore, using limiter stages before the detector is necessary.
View (A) of figure 3-11 shows a typical Foster-Seeley discriminator. The collector circuit of the preceding limiter/amplifier circuit (Q1) is shown. The limiter/amplifier circuit is a special amplifier circuit which limits the amplitude of the signal. This limiting keeps interfering noise low by removing
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excessive amplitude variations from signals. The collector circuit tank consists of C1 and L1. C2 and L2 form the secondary tank circuit. Both tank circuits are tuned to the center frequency of the incoming fm signal. Choke L3 is the dc return path for diode rectifiers CR1 and CR2. R1 and R2 are not always necessary but are usually used when the back (reverse bias) resistance of the two diodes is different. Resistors R3 and R4 are the load resistors and are bypassed by C3 and C4 to remove rf. C5 is the output coupling capacitor.
CIRCUIT OPERATION AT RESONANCE.—The operation of the Foster-Seeley discriminator can best be explained using vector diagrams [figure 3-11, view (B)] that show phase relationships between the voltages and currents in the circuit. Let's look at the phase relationships when the input frequency is equal to the center frequency of the resonant tank circuit.
The input signal applied to the primary tank circuit is shown as vector ep. Since coupling capacitor C8 has negligible reactance at the input frequency, rf choke L3 is effectively in parallel with the primary tank circuit. Also, because L3 is effectively in parallel with the primary tank circuit, input voltage ep also appears across L3. With voltage ep applied to the primary of T1, a voltage is induced in the secondary which causes current to flow in the secondary tank circuit. When the input frequency is equal to the center frequency, the tank is at resonance and acts resistive. Current and voltage are in phase in a resistance circuit, as shown by is and ep. The current flowing in the tank causes voltage drops across each half of the balanced secondary winding of transformer T1. These voltage drops are of equal amplitude and opposite
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polarity with respect to the center tap of the winding. Because the winding is inductive, the voltage across it is 90 degrees out of phase with the current through it. Because of the center-tap arrangement, the voltages at each end of the secondary winding of T1 are 180 degrees out of phase and are shown as e1 and e2 on the vector diagram.
The voltage applied to the anode of CR1 is the vector sum of voltages ep and e1, shown as e 3 on the diagram. Likewise, the voltage applied to the anode of CR2 is the vector sum of voltages ep and e 2, shown as e4 on the diagram. At resonance e3 and e4 are equal, as shown by vectors of the same length. Equal anode voltages on diodes CR1 and CR2 produce equal currents and, with equal load resistors, equal and opposite voltages will be developed across R3 and R4. The output is taken across R3 and R4 and will be 0 at resonance since these voltages are equal and of appositive polarity.
The diodes conduct on opposite half cycles of the input waveform and produce a series of dc pulses at the rf rate. This rf ripple is filtered out by capacitors C3 and C4.
OPERATION ABOVE RESONANCE.—A phase shift occurs when an input frequency higher than the center frequency is applied to the discriminator circuit and the current and voltage phase relationships change. When a series-tuned circuit operates at a frequency above resonance, the inductive reactance of the coil increases and the capacitive reactance of the capacitor decreases. Above resonance the tank circuit acts like an inductor. Secondary current lags the primary tank voltage, ep. Notice that secondary voltages e
1 and e2 are still 180 degrees out of phase with the current (iS) that produces them. The change to a lagging secondary current rotates the vectors in a clockwise direction. This causes el to become more in phase with ep while e2 is shifted further out of phase with ep. The vector sum of ep and e2 is less than that of ep and e1. Above the center frequency, diode CR1 conducts more than diode CR2. Because of this heavier conduction, the voltage developed across R3 is greater than the voltage developed across R4; the output voltage is positive.
OPERATION BELOW RESONANCE.—When the input frequency is lower than the center frequency, the current and voltage phase relationships change. When the tuned circuit is operated at a frequency lower than resonance, the capacitive reactance increases and the inductive reactance decreases. Below resonance the tank acts like a capacitor and the secondary current leads primary tank voltage ep. This change to a leading secondary current rotates the vectors in a counterclockwise direction. From the vector diagram you should see that e2 is brought nearer in phase with ep, while el is shifted further out of phase with ep. The vector sum of ep and e2 is larger than that of e and e1. Diode CR2 conducts more than diode CR1 below the center frequency. The voltage drop across R4 is larger than that across R3 and the output across both is negative.
These voltage outputs can be plotted to show the response curve of the discriminator discussed earlier (figure 3-10). When weak AM signals (too small in amplitude to reach the circuit limiting level) pass through the limiter stages, they can appear in the output. These unwanted amplitude variations will cause primary voltage ep [view (A) of figure 3-11] to fluctuate with the modulation and to induce a similar voltage in the secondary of T1. Since the diodes are connected as half-wave rectifiers, these small AM signals will be detected as they would be in a diode detector and will appear in the output. This unwanted AM interference is cancelled out in the ratio detector (to be studied next in this chapter) and is the main disadvantage of the Foster-Seeley circuit.
Q-23. What type of tank circuit is used in the Foster-Seeley discriminator? Q-24. What is the purpose of CR1 and CR2 in the Foster-Seeley discriminator? Q-25. What type of impedance does the tank circuit have above resonance?
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The RATIO DETECTOR uses a double-tuned transformer to convert the instantaneous frequency variations of the fm input signal to instantaneous amplitude variations. These amplitude variations are then rectified to provide a dc output voltage which varies in amplitude and polarity with the input signal frequency. This detector demodulates fm signals and suppresses amplitude noise without the need of limiter stages.
Figure 3-12 shows a typical ratio detector. The input tank capacitor (C1) and the primary of transformer T1 (L1) are tuned to the center frequency of the fm signal to be demodulated. The secondary winding of T1 (L2) and capacitor C2 also form a tank circuit tuned to the center frequency. Tertiary (third) winding L3 provides additional inductive coupling which reduces the loading effect of the secondary on the primary circuit. Diodes CR1 and CR2 rectify the signal from the secondary tank. Capacitor C5 and resistors R1 and R2 set the operating level of the detector. Capacitors C3 and C4 determine the amplitude and polarity of the output. Resistor R3 limits the peak diode current and furnishes a dc return path for the rectified signal. The output of the detector is taken from the common connection between C3 and C4. Resistor RL is the load resistor. R5, C6, and C7 form a low-pass filter to the output.
This circuit operates on the same principles of phase shifting as did the Foster-Seeley discriminator. In that discussion, vector diagrams were used to illustrate the voltage amplitudes and polarities for conditions at resonance, above resonance, and below resonance. The same vector diagrams apply to the ratio detector but will not be discussed here. Instead, you will study the resulting current flows and polarities on simplified schematic diagrams of the detector circuit.
OPERATION AT RESONANCE.—When the input voltage ep is applied to the primary in figure 3-12 it also appears across L3 because, by inductive coupling, it is effectively connected in parallel with the primary tank circuit. At the same time, a voltage is induced in the secondary winding and causes current to flow around the secondary tank circuit. At resonance the tank acts like a resistive circuit; that is,
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the tank current is in phase with the primary voltage ep. The current flowing in the tank circuit causes voltages e1 and e2 to be developed in the secondary winding of T1. These voltages are of equal magnitude and of opposite polarity with respect to the center tap of the winding. Since the winding is inductive, the voltage drop across it is 90 degrees out of phase with the current through it.
Figure 3-13 is a simplified schematic diagram of a ratio detector at resonance. The voltage applied to the cathode of CR1 is the vector sum of e1 and ep. Likewise, the voltage applied to the anode of CR2 is the vector sum of e2 and ep. No phase shift occurs at resonance and both voltages are equal. Both diodes conduct equally. This equal current flow causes the same voltage drop across both R1 and R2. C3 and C4 will charge to equal voltages with opposite polarities. Let’s assume that the voltages across C3 and C4 are equal in amplitude (5 volts) and of opposite polarity and the total charge across C5 is 10 volts. R1 and R2 will each have 5 volts dropped across them because they are of equal values. The output is taken between points A and B. To find the output voltage, you algebraically add the voltages between points A and B (loop ACB or ADB). Point A to point D is - 5 volts. Point D to point B is + 5 volts. Their algebraic sum is 0 volts and the output voltage is 0 at resonance. If the voltages on branch ACB were figured, the same output would be found because the circuit branches are in parallel.
When the input signal reverses polarity, the secondary voltage across L2 also reverses. The diodes will be reverse biased and no current will flow. Meanwhile, C5 retains most of its charge because of the long time constant offered in combination with R1 and R2. This slow discharge helps to maintain the output.
OPERATION ABOVE RESONANCE.—When a tuned circuit (figure 3-14) operates at a frequency higher than resonance, the tank is inductive. The secondary current i
lags the primary voltage ep. Secondary voltage e1 is nearer in phase with primary voltage e, while e2 is shifted further out of phase with ep. The vector sum of e1 and ep is larger than that of e2 and ep. Therefore, the voltage applied to the cathode of CR1 is greater than the voltage applied to the anode of CR2 above resonance.
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Assume that the voltages developed above resonance are such that the higher voltage on the cathode of CR1 causes C3 to charge to 8 volts. The lower voltage on the anode of CR2 causes C4 to charge to 2 volts. Capacitor C5 remains charged to the sum of these two voltages, 10 volts. Again, by adding the voltages in loop ACB or ADB between points A and B, you can find the output voltage. Point A to point D equals -2 volts. Point D to point B equals +5 volts. Their algebraic sum, and the output, equals +3 volts when tuned above resonance. During the negative half cycle of the input signal, the diodes are reverse biased and C5 helps maintain a constant output.
OPERATION BELOW RESONANCE.—When a tuned circuit operates below resonance (figure 3-15), it is capacitive. Secondary current is leads the primary voltage ep and secondary voltage e2 is nearer in phase with primary voltage ep. The vector sum of e2 and ep is larger than the sum of e1 and ep. The voltage applied to the anode of CR2 becomes greater than the voltage applied to the cathode of CR1 below resonance.
Assume that the voltages developed below resonance are such that the higher voltage on the anode of CR2 causes C4 to charge to 8 volts. The lower voltage on the cathode of CR1 causes C3 to charge to 2 volts. Capacitor C5 remains charged to the sum of these two voltages, 10 volts. The output voltage equals - 8 volts plus +5 volts, or - 3 volts, when tuned below resonance. During the negative half cycle of the input signal, the diodes are reverse biased and C5 helps maintain a constant output.
The ratio detector is not affected by amplitude variations on the fm wave. The output of the detector adjusts itself automatically to the average amplitude of the input signal. C5 charges to the sum of the voltages across R1 and R2 and, because of its time constant, tends to filter out any noise impulses. Before C5 can charge or discharge to the higher or lower potential, the noise disappears. The difference in charge across C5 is so slight that it is not discernible in the output. Ratio detectors can operate with as little as 100 millivolts of input. This is much lower than that required for limiter saturation and less gain is required from preceding stages.
Q-26. What is the primary advantage of a ratio detector? Q-27. What is the purpose of C5 in figure 3-12?
An fm demodulator employing a completely different detection principle is the GATED-BEAM DETECTOR (sometimes referred to as the QUADRATURE DETECTOR). A simplified diagram of a
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gated-beam detector is shown in figure 3-16. It uses a gated-beam tube to limit, detect, and amplify the received fm signal. The output voltage is 0 when the input frequency is equal to the center frequency. When the input frequency rises above the center frequency, the output voltage goes positive. When the input frequency drops below the center frequency, the output voltage goes negative.
The gated-beam detector employs a specially designed gated-beam tube. The elements of this tube are shown in figure 3-17. The focus electrode forms a shield around the tube cathode except for a narrow slot through which the electron beam flows. The beam of electrons flows toward the limiter grid which acts like a gate. When the gate is open, the electron beam flows through to the next grid. When closed, the gate completely stops the beam.
After the electron beam passes the limiter grid, the screen grid refocuses the beam toward the quadrature grid. The quadrature grid acts much the same as the limiter grid; it either opens or closes the passage for electrons. These two grids act similar to an AND gate in digital devices; both gates must be open for the passage of electrons to the plate. Either grid can cut off plate current. AND gates were presented in NEETS, Module 13, Introduction to Number Systems, Boolean Algebra, and Logic Circuits.
Look again at the circuit in figure 3-16. With no signal applied to the limiter grid (3), the tube conducts. The electron beam moving near the quadrature grid (5) induces a current into the grid which develops a voltage across the high-Q tank circuit (L3 and C3). C3 charges until it becomes sufficiently
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negative to cut off the current flow. L3 tends to keep the current moving and, as its field collapses, discharges C3. When C3 discharges sufficiently, the quadrature grid becomes positive, grid current flows, and the cycle repeats itself. This tank circuit (L3 and C3) is tuned to the center frequency of the received fm signal so that it will oscillate at that frequency.
The waveforms for the circuit are shown in figure 3-18. View (A) is the fm input signal. The limiter-grid gate action creates a wave shape like view (B) because the tube is either cut off or saturated very quickly by the input wave. Note that this is a square wave and is the current waveform passing the limiter grid.
At the quadrature grid the voltage across C3 lags the current which produces it [view (C)]. The result is a series of pulses, shown in view (D), appearing on the quadrature grid at the center frequency, but lagging the limiter-grid voltage by 90 degrees. Because the quadrature grid has the same conduction and cutoff levels as the limiter grid, the resultant current waveform will be transformed into a square wave.
Both the limiter and quadrature grids must be positive at the same time to have plate current. You can see how much conduction time occurs for each cycle of the input by overlaying the current waveforms in views (B) and (D), as shown in view (E). The times when both grids are positive are shown by the shaded area of view (E). These plate current pulses are shown for operation at resonance in view (F).
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Now consider what happens with a deviation in frequency at the input. If the frequency increases, the frequency across the quadrature tank also increases. Above resonance, the tank appears capacitive to the induced current; voltage then lags the applied voltage by more than 90 degrees, as shown in view (G). Note in view (H) that the two grid signals have moved more out of phase and the average plate current level has decreased.
As the input frequency decreases, the opposite action takes place. The two grid signals move more in phase, as shown in view (I), and the average plate current increases, as shown in view (J).
View (K) shows the resultant plate-current pulses when an fm signal is applied to a gated-beam detector. Plate load resistor R4 and capacitor C6 form an integrating network which filters these pulses to form the sine-wave output.
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The primary advantage of the gated-beam detector lies in its extreme simplicity. It employs only one tube, yet provides a very effective limiter with linear detection. It requires relatively few components and is very easily adjusted.
There are more than the three types of fm demodulators presented in this chapter. However, these are representative of the types with which you will be working. The principles involved in their operation are similar to the other types. You will now briefly study PHASE DEMODULATION which uses the same basic circuitry as fm demodulators.
Q-28. What circuit functions does the tube in a gated-beam detector serve? Q-29. What condition must exist on both the limiter and quadrature grids for current to flow in a
gated-beam detector?
Q-30. Name two advantages of the gated-beam detector.
In phase modulation (pm) the intelligence is contained in the amount and rate of phase shift in a carrier wave. You should recall from your study of pm that there is an incidental shift in frequency as the phase of the carrier is shifted. Because of this incidental frequency shift, fm demodulators, such as the Foster-Seeley discriminator and the ratio detector, can also be used to demodulate phase-shift signals.
Another circuit that may be used is the gated-beam (quadrature) detector. Remember that the fm phase detector output was determined by the phase of the signals present at the grids. A QUADRATURE DETECTOR FOR PHASE DEMODULATION works in the same manner.
A basic schematic is shown in figure 3-19. The quadrature-grid signal is excited by a reference from the transmitter. This may be a sample of the unmodulated master oscillator providing a phase reference for the detector.
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The modulated waveform is applied to the limiter grid. Gating action in the tube will occur as the phase shifts between the input waveform and the reference. The combined output current from the gated- beam tube will be a series of current pulses. These pulses will vary in width as shown in figure 3-20. The width of these pulses will vary in accordance with the phase difference between the carrier and the modulated wave.
Q-31. Where is the intelligence contained in a phase-modulated signal? Q-32. Why can phase-modulated signals be detected by fm detectors? Q-33. How is a quadrature detector changed when used for phase demodulation?
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Pulse modulation is used in radar circuits as well as communications circuits, as discussed in chapter 2. A pulse-modulated signal in radar may be detected by a simple circuit that detects the presence of rf energy. Circuits that are capable of this were covered in this chapter in the cw detection discussion; therefore, the information will not be repeated here. A RADAR DETECTOR, in its simplest form, must be capable of producing an output when rf energy (reflected from a target) is present at its input.
In COMMUNICATIONS PULSE DETECTORS the modulated waveform must be restored to its original form. In this chapter you will study three basic methods of pulse demodulation: PEAK, LOW-PASS FILTER, and CONVERSION.
Peak detection uses the amplitude of a pulse-amplitude modulated (pam) signal or the duration of a pulse-duration modulated (pdm) signal to charge a holding capacitor and restore the original waveform. This demodulated waveform will contain some distortion because the output wave is not a pure sine wave. However, this distortion is not serious enough to prevent the use of peak detection.
Peak detection is used to detect pam. Figure 3-21 includes a simplified circuit [view (A)] for this demodulator and its waveforms [views (B) and (C)]. CR1 is the input diode which allows capacitor C1 to charge to the peak value of the pam input pulse. Pam input pulses are shown in view (B). CR1 is reverse biased between input pulses to isolate the detector circuit from the input. CR2 and CR3 are biased so that they are normally nonconducting. The discharge path for the capacitor is through the resistor (R1). These components are chosen so that their time constant is at least 10 times the interpulse period (time between pulses). This maintains the charge on C1 between pulses by allowing only a small discharge before the next pulse is applied. The capacitor is discharged just prior to each input pulse to allow the output voltage to follow the peak value of the input pulses. This discharge is through CR2 and CR3. These diodes are turned on by a negative pulse from a source that is time-synchronous with the timing-pulse train at the transmitter. Diode CR3 ensures that the output voltage is near 0 during this discharge period. View (C) shows the output wave shape from this circuit. The peaks of the output signal follow very closely the original modulating wave, as shown by the dotted line. With additional filtering this stepped waveform closely approximates its original shape.
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The peak detector circuit may also be used for pdm. To detect pdm, you must modify view (A) of figure 3-21 so that the time constant for charging C1 through CR1 is at least 10 times the maximum received pulse width. This may be done by adding a resistor in series with the cathode or anode circuit of CR1. The amplitude of the voltage to which C1 charges, before being discharged by the negative pulse, will be directly proportional to the input pulse width. A longer pulse width allows C1 to charge to a higher potential than a short pulse. This charge is held, because of the long time constant of R1 and C1, until the discharge pulse is applied to diodes CR2 and CR3 just prior to the next incoming pulse. These charges across C1 result in a wave shape similar to the output shown for pam detection in view (C) of figure 3-21.
Q-34. In its simplest form, what functions must a radar detector be capable of performing? Q-35. What characteristic of a pulse does a peak detector sample? Q-36. What is the time constant of the resistor and capacitor in a peak detector for pam? Q-37. How can a peak detector for pam be modified to detect pdm?
Another method of demodulating pdm is by the use of a low-pass filter. If the voltage of a pulse waveform is averaged over both the pulse and no-pulse time, average voltage is the result. Since the amplitude of pdm pulses is constant, average voltage is directly proportional to pulse width. The pulse width varies with the modulation (intelligence) in pdm. Because the average value of the pulse train varies in accordance with the modulation, the intelligence may be extracted by passing the width- modulated pulses through a low-pass filter. The components of such a filter must be selected so that the filter passes only the desired modulation frequencies. As the varying-width pulses are applied to the low-
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pass filter, the average voltage across the filter will vary in the same way as the original modulating voltage. This varying voltage will closely approximate the original modulating voltage.
Pulse-position modulation (ppm), pulse-frequency modulation (pfm), and pulse-code modulation (pcm) are most easily demodulated by first converting them to either pdm or pam. After conversion these pulses are demodulated using either peak detection or a low-pass filter. This conversion may be done in many ways, but your study will be limited to the simpler methods.